1. Technical Field
The disclosed embodiments relate to low noise amplifiers.
2. Background Information
The first amplification stage in a radio receiver such as a receiver of a cellular telephone is generally an amplifier circuit called a Low Noise Amplifier (LNA). The LNA reduces the noise contributions of following stages and sets the lowest achievable noise level of the overall receiver. The LNA is therefore generally designed to have high gain to minimize the amount of noise introduced without introducing unacceptably large amounts of distortion. If a sinusoidal input signal of a pure single frequency is supplied to the input of a linear amplifier, then the amplifier will output an amplified version of the input signal. The output signal will have only a single frequency, and this frequency will be the frequency of the input signal. If, however, the same sinusoidal input signal is supplied to the input of an amplifier that exhibits an amount of non-linearity, then the amplifier will output an amplified version of the input signal at the frequency of the input signal, but the amplifier will also output one or more other signals of other frequencies. These other signals are referred to as “distortion”. The interactions between the input signal (or multiple input signals) and the particular non-linearities of the amplifier can be complex, and the type of distortion can also be complex and depends on many different characteristics of the amplifier and the input signal.
More particularly, the transconductance of a non-linear amplifier is sometimes described using an equation of the form below:y=go+g1x+g2x2+g3x3  (Equ. 1)In Equation 1, the x denotes an input signal and the y denotes a resulting output signal. The terms g1x and g2x2 and g3x3 are referred to as the first-order term (or “linear term”), the second-order term, and the third-order term, respectively. If x (the input signal) is a voltage and y (the output signal) is a current, then g1 is referred to as the “linear transconductance coefficient” while the coefficients g2 and g3 are referred to as the second order transconductance coefficient, and the third order transconductance coefficient, respectively.
As output power of an LNA is increased, the magnitude of generated distortion increases faster than does the magnitude of the desired signal. At some output power, the magnitude of distortion is equal to the magnitude of the desired signal. This output power where the magnitude of third order distortion is equal to the output power of the desired signal is referred to as the third-order intercept point IIP3. To a first approximation, IIP3 is given by Equation 2 below if the second order contribution to IIP3 is neglected:IIP3=√{square root over ((4/3)(g1/g3))}{square root over ((4/3)(g1/g3))}  (Equ. 2)
The receiver in a cellular telephone may be used to receive a signal in a condition in which there are undesired signals in addition to the desired signal to be received. These undesired signals are called jammers and they can have very different natures. Adjacent channel signals and transmitter signals are just some examples of jammers. Jammers can be discrete tones or can have a given bandwidth. For analysis purposes, a non-discrete signal can be modeled as two sine wave signals, each having a different frequency, where the difference in frequencies of the two sine wave signals is the bandwidth of the non-discrete signal.
Consider a situation in which the desired signal to be received has a frequency of 1 GHz. If a first adjacent channel receive jammer has a frequency ω1 of 1.001 GHz, and a second adjacent channel receive jammer has a frequency ω2 of 1.002 GHz, and if the sum of these two jammer signals is supplied as variable x into Equation 1 above, then the resulting y signal will, due to the squaring and cubing of terms, have many components of many frequencies. Due to the third order term and associated cubing of the sum of input signals, there will be one component of the output y that has a frequency of (2ω1−ω2). This component is therefore at the same 1 GHz frequency as the desired signal. Because this distortion component and the desired signal have the same 1 GHz frequency, the desired signal cannot be separated from the distortion component by filtering. A receiver that is more linear is therefore desired in order to reduce the magnitude of this distortion component. This distortion component is sometimes referred to as the third order “intermodulation distortion”.
In addition, in a cellular telephone that operates in accordance with a Code Division Multiple Access (CDMA) standard such as CDMA2000, the cellular telephone has a transmitter that may be transmitting at the same time that the receiver of the cellular telephone is receiving. Although the transmitted signals are transmitted in a different frequency band than the desired signal that is being received, the transmitted signals can be strong and are output from the cellular telephone transmitter in close proximity to the highly sensitive receiver of the cellular telephone. Accordingly, a substantial amount of the transmitted signals may leak back into the receiver and cause distortion problems. The transmitted signals are transmitted in a band, so they may be modeled as described above as two signals that have different frequencies, where the difference in frequencies of the two signals is the width of the channel.
Then, in addition, there is the signal that is to be received. This signal is referred to as the desired signal. Consider a situation in which the desired signal to be received has a receive frequency ω3 of 1 GHz. If a first transmit jammer is at a frequency ω1 of 900.0 MHz, and a second transmit jammer is at a frequency ω2 of 900.4 MHz, and if the sum of the two jammers signals and the desired signal is supplied as variable x into Equation 1 above, then the resulting y signal will, due to the squaring and cubing of terms, have many components of many frequencies. Due to the third order term and the resulting cubing, it so happens there will be one component of the output y that has a frequency of ω3−(ω2−ω1). This frequency in this example is 1.0006 GHz, and is therefore in the one megahertz wide receive band centered at 1 GHz. This component is sometimes referred to as the “triple beat” distortion component, or the third order “cross-modulation” component. Because the cross-modulation component is in the receive band, it typically cannot be separated from the desired signal by filtering. A receiver that is more linear is therefore also desired in order to reduce the magnitude of this cross-modulation distortion component.
In addition to introducing no more than an acceptable amount of distortion, the amplifier should introduce a minimal amount of noise. Thermal noise is due to random motion of electrons and atoms within the resistive component of any impedance such as the resistive component of the semiconductor material of which transistors are made. All amplifiers utilizing transistors therefore introduce noise. This noise is naturally occurring and inherent in the amplifier. The introduction of noise into the output of the LNA cannot be eliminated, but poor design can result in the LNA adding noise more than is necessary and amplifying pre-existing noise more than necessary. The noise output by the LNA then flows through the remainder of the receiver. An LNA is therefore designed to reduce and minimize the amount of noise the LNA outputs.
Several different techniques and circuit topologies are conventionally applied to realize LNAs that exhibit low noise and distortion performance. These techniques include techniques referred to as feed-back cancellation, pre-distortion cancellation, feed-forward cancellation, and post-distortion cancellation. Three specific examples of post-distortion cancellation techniques are of interest here and are referred to as the Active Post-Distortion (APD) technique, the Derivative Super-position (DS) technique, and the Modified Derivative Superposition (MDS) technique.
FIG. 1 (Prior Art) is a circuit diagram of a differential LNA 1 that utilizes the Active Post-Distortion technique. This technique involves the use of four field effect transistors (FETs) biased in the saturation region. FETs 2 and 3 are referred to as the main FETs. FETs 4 and 5 are referred to as the cancel FETs. One pair of main FET and cancel FET operates as follows. Main FET 2 (which defines the gain and noise figure of the circuit) amplifies an input signal on input lead 5. An amplified version of the input signal is generated onto node 6. Because main FET 2 is configured as a common source amplifier, the amplified signal has a phase shift of approximately 180 degrees with respect to the input signal on input lead 5. Third order distortion components are also present in the signal on node 6 along with the desired amplified version of the input signal. The phase-shifted signal on node 6 is applied to the input of cancel FET 4. Cancel FET 4 is biased in the saturation region, but nonetheless is has a significantly non-linear amplifying characteristic. Cancel FET 4 is designed to be a lousy amplifier in that it generates a substantial amount of third order distortion but supplies only a small amount of the desired signal, in amplified form, onto its drain. The magnitude of the distortion signal output by FET 4 is set to be equal in magnitude to the distortion signal output by FET 2 onto node 6. Because cancel FET 4 is biased in the saturation region, both the distortion it outputs as well as the amplified desired signal it outputs are 180 degrees out of phase with respect to the third order distortion components on node 6. The current signals output from main FET 2 and cancel FET 4 are summed at node 7. This summing results in cancellation of the third order distortion in both signals. Unfortunately, besides canceling the unwanted third order distortion, this technique also results in some cancellation of the desired signal because the amplified versions of the desired input signal that are output by FETs 2 and 4 are in phase with each other. The gain of the LNA is therefore degraded. See Published U.S. Patent Application No. 2007/0229154, published Oct. 4, 2007, for further details on the differential LNA of FIG. 1.
FIG. 2 (Prior Art) is a circuit diagram of a single-ended LNA 10 that utilizes the Derivative Super-position (DS) technique. This example is a single ended circuit, as opposed to the differential circuit of the example of FIG. 1. In the DS circuit of FIG. 2, two FETs 11 and 12 are used. FET 12 is referred to as the main FET and it is biased in the saturation region. FET 11 is referred to as cancel FET and it is biased in the sub-threshold region. When the transconductance equation for a FET amplifier biased in the saturation region is compared to the transconductance equation for a FET amplifier biased in the sub-threshold region, it is recognized that the signs of the third order coefficients of the transconductance equations of the two transistors are opposite one another. The signs of the first order coefficients, however, are not opposite one another. This means that biasing a transistor in the sub-threshold region results in a shift in the phase of the third order distortion signal it outputs as compared to a transistor biased in the saturation region, whereas the phase of the desired signal as output by the sub-threshold biased transistor is not phase shifted as compared to the transistor biased in the saturation region. The currents output by FETS 11 and 12 are therefore summed on node 13, resulting in the distortion signal output by the cancel FET 11 canceling the third order distortion output by main FET 12. The signal that cancel FET 11 outputs that is of the frequency of the desired input signal is not, however, 180 degrees out of phase with respect to the amplified version of the desired signal as output by main FET 12, and consequently a portion of the desired signal on node 13 is not cancelled as in the APS example of FIG. 1.
Employing the DS technique of FIG. 2, however, has a problem. The source degeneration inductance 14 creates a feedback path, which allows the second order transconductance coefficient to contribute to the third order distortion. As a result, the DS technique does not significantly increase the third-order intercept point IIP3. In the DS technique, the second order contribution of third order distortion results in an undesirably low IIP3.
FIG. 3 (Prior Art) is a diagram of an LNA 15 employing the Modified Derivative Superposition (MDS) technique. Rather then scaling and rotating the second order transconductance coefficient g2MAIN contribution to third-order intermodulation distortion as in the case of the DS technique, the MDS technique changes the magnitude and phase of the third order transconductance coefficient g3CANCEL contribution to the third order intermodulation distortion relative to third order transconductance coefficient g3MAIN contribution to the third order intermodulation distortion, such that their sum (g3CANCEL and g3MAIN contributions) is out-of-phase with the second order coefficient g2MAIN contribution to the third order intermodulation distortion. A purpose of connecting the source of cancel FET 16 to the common node of the two inductors 17 and 18 is to change the magnitude and phase of the g3CANCEL contribution relative to the g2MAIN and g3MAIN contributions of main FET 19. The MDS LNA 15 of FIG. 3 therefore has an improved IIP3 as compared to the IIP3 of the DS LNA 10 of FIG. 2. It is to be understood that the description of the phase relationships and transistor operations set forth above are simplifications. They are presented here for instructional purposes. See Published U.S. Patent Application No. 2005/0176399, published Aug. 11, 2005, for a more detailed explanation of the operation of an LNA that employs the Modified Derivative Superposition (MDS) technique.
Unfortunately, an amplifier that employs a sub-threshold-biased FET is generally a noisy amplifier as compared to an amplifier that employs a FET that is biased in the saturation region. In the MDS LNA circuit 15 of FIG. 3, cancel FET 16 introduces an undesirable amount of noise into the LNA output. Moreover, the gate of cancel FET 16 is coupled to the input lead 20 of LNA 15, resulting in LNA 15 having an undesirably large input capacitance.